Mam sprawny zasilacz 18V 1.1A od drukarki HP (chyba od DJ 840c). Zasilacz ma układ Astec AS3843 (Link). Chciałbym przerobić ten zasilacz - w opcji pierwszej - na napięcie 12V i najlepiej 1.25A lub ciut więcej amperów. Opcja druga - po postu na 12 V 1.1A. Jak wykonać takie przeróbki? Zdjęcia płytki: http://obrazki.elektroda.pl/7392603700_1351276886_thumb.jpg http://obrazki.elektroda.pl/4709190900_1351278531_thumb.jpg Na płytce jest transoptor SFH617A - element IC2 Link W załącznikach są dane elementów AS3843 oraz SFH617A.
AS384x
Current Mode Controller
SEMICONDUCTOR
Features
Description
•
2.5 V bandgap reference trimmed to
1.0% and temperature-compensated
•
Standard temperature range extended
to 105°C
•
AS3842/3 oscillations trimmed for
precision duty cycle clamp
The AS3842 family of control ICs provide pin-for-pin replacement of the
industry standard UC3842 series of devices. The devices are
redesigned to provide significantly improved tolerances in power supply
manufacturing. The 2.5 V reference has been trimmed to 1.0%
tolerance. The oscillator discharge current is trimmed to provide guaranteed duty cycle clamping rather than specified discharge current. The
circuit is more completely specified to guarantee all parameters impacting power supply manufacturing tolerances.
•
AS3844/5 have exact 50% max duty
cycle clamp
•
Advanced oscillator design simplifies
synchronization
•
Improved specs on UVLO and
hysteresis provide more predictable
start-up and shutdown
•
Improved 5 V regulator provides better
AC noise immunity
•
Guaranteed performance with current
sense pulled below ground
Pin Configuration —
In addition, the oscillator and flip-flop sections have been enhanced to
provide additional performance. The RT/CT pin now doubles as a synchronization input that can be easily driven from open collector/open
drain logic outputs. This sync input is a high impedance input and can
easily be used for externally clocked systems. The new flip-flop topology allows the duty cycle on the AS3844/5 to be guaranteed between
49 and 50%. The AS3843/5 requires less than 0.5 mA of start-up current over the full temperature range.
Top view
PDIP (N)
8L SOIC (8D)
COMP
1
8
VREG
VFB
2
7
VCC
ISENSE
3
6
RT/CT
4
5
14L SOIC (14D)
COMP
1
8
VREG
COMP
1
14 VREG
VFB
2
7
VCC
NC
2
13 NC
OUT
ISENSE
3
6
OUT
VFB
3
12 VCC
GND
RT/CT
4
5
GND
NC
4
11 VC
ISENSE
5
10 OUT
NC
6
9
PWR GND
RT/CT
7
8
GND
Ordering Information
AS384X
8D
N
Circuit Type:
Current Mode Controller (See Table A)
Table A
Package Style
8D = 8 Pin Plastic SOIC
14D = 14 Pin Plastic SOIC
N = 8 Pin Plastic DIP
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Model
AS3842
AS3843
AS3844
AS3845
61
Packaging Option:
N = Tape and Reel (13 " Reel Dia)
T = Tube
VCC(min)
10
7.6
10
7.6
VCC(on)
16
8.4
16
8.4
Duty Cycle
Typ.
97%
97%
49.5%
49.5%
ICC
0.5 mA
0.3 mA
0.5 mA
0.3 mA
AS384x
Current Mode Controller
Functional Block Diagram
(5.0 V)
(5.0 V)
5V
REGULATOR
1
COMP
8
VREG
(2.5 V)
REF OK
+
2
VFB
(4 V)
–
2R
7
VCC
UVLO
ERROR AMP
(1.0 V)
(6 V)
R
PWM
COMPARATOR
3
CURRENT
SENSE
FF
–
S
+
PWM LOGIC
6
OUTPUT
R
(5 V)
–
(3.0 V)
CLK ÷ 2 [3844/45]
+
–
(1.3 V)
S
–
(0.6 V)
CLK [3842/43]
FF
+
4
RT/CT
FF
5
GND
R
T
OSCILLATOR
+
OVER
TEMPERATURE
Figure 1. Block Diagram of the AS3842/3/4/5
Pin Function Description
Pin Number Function
Description
1
COMP
2
VFB
3
Current
Sense
A voltage proportional to inductor current is connected to the input. The PWM uses
this information to terminate the gate drive of the output.
4
RT/CT
Oscillator frequency and maximum output duty cycle are set by connecting a resistor
(RT) to VREG and a capacitor (CT) to ground. Pulling this pin to ground or to VREG will
accomplish a synchronization function.
5
GND
Circuit common ground, power ground, and IC substrate.
6
Output
7
VCC
Positive supply voltage for the IC.
8
VREG
This 5 V regulated output provides charging current for the capacitor CT through the
resistor RT.
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This pin is the error amplifier output. Typically used to provide loop compensation to
maintain VFB at 2.5 V.
Inverting input of the error amplifier. The non-inverting input is a trimmed 2.5 V
bandgap reference.
This output is designed to directly drive a power MOSFET switch. This output can sink
or source peak currents up to 1A. The output for the AS3844/5 switches at one-half
the oscillator frequency.
62
AS384x
Current Mode Controller
Electrical Characteristics
(cont’d)
Electrical characteristics are guaranteed over full junction temperature range (0 to 105°C). Ambient temperature must be derated
based on power dissipation and package thermal characteristics. The conditions are: VCC = 15 V, RT = 10 kΩ, and CT = 3.3 nF, unless
otherwise stated. To override UVLO, VCC should be raised above 17 V prior to test.
Parameter
Symbol
Test Condition
Min.
Typ.
Max.
Unit
Dmax
3842/3
94
97
100
%
Minimum Duty Cycle
Dmin
3842/3
Maximum Duty Cycle
Dmax
3844/5
0
%
49
49.5
50
%
Minimum Duty Cycle
Dmin
3844/5
0
%
ICC
3842/4, VFB = VSENSE = 0 V, VCC = 14 V
3843/5, VFB = VSENSE = 0 V, VCC = 7 V
0.5
1.0
mA
0.3
0.5
mA
ICC = 25 mA
30
9
17
mA
PWM
Maximum Duty Cycle
Supply Current
Start-up Current
Operating Supply Current
ICC
VCC Zener Voltage
VZ
V
Notes:
1. This parameter is not 100% tested in production.
2. Parameter measured at trip point of PWM latch.
3. Transfer gain is the relationship between current sense input and corresponding error amplifier output at the PWM latch trip point
and is mathematically expressed as follows:
A=
∆ I COMP
∆VSENSE
; – 0.2 ≤ V SENSE ≤ 0.8 V
4. At the over-temperature threshold, TOT, the oscillator is disabled. The 5 V reference and the PWM stages, including the PWM latch,
remain powered.
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66
Current Mode Controller
AS384x
Typical Performance Curves
Supply Current vs Supply Voltage
Output Voltage vs Supply Voltage
20
15
10
10
0
AS3843/5
5
AS3842/4
AS3843/5
5
15
0
0
5
20
25
10
15
VCC – Supply Voltage (V)
30
35
0
5
15
20
25
30
Figure 3
Regulator Output Voltage vs Ambient Temperature
Regulator Short Circuit Current vs Ambient Temperature
5.04
160
5.02
IREG – Regulator Short Circuit (mA)
140
5.00
VREG – Regulator Output (V)
10
VCC – Supply Voltage (V)
Figure 2
4.98
4.96
4.94
4.92
4.90
–60
AS3842/4
VOUT – Output Voltage (V)
25
20
ICC – Supply Current (mA)
25
–30
0
30
60
90
TA – Ambient Temperature (°C)
120
100
80
60
40
–60
150
Figure 4
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120
–30
0
30
60
90
TA – Ambient Temperature (°C)
Figure 5
67
120
150
AS384x
Current Mode Controller
Typical Performance Curves
Regulator Load Regulation
Maximum Duty Cycle vs Timing Resistor
100
–4
Maximum Duty Cycle (%)
∆VREG – Regulator Voltage Change (mV)
0
–8
–12
25°C
150°C
–55°C
–16
80
60
40
–20
20
0.3
–24
0
20
40
60
80
100
120
140
1
3
RT – Timing Register (kΩ)
ISC – Regulator Source Current (mA)
Figure 6
Figure 7
Maximum Duty Cycle Temperature Stability
Timing Capacitor vs Oscillator Frequency
100
100
Maximum Duty Cycle (%)
10
RT = 2.2 kΩ
RT = 4.7 kΩ
1
RT = 10 kΩ
90
RT = 680 Ω
CT – Timing Capacitor (nF)
10
RT = 1 kΩ
RT = 2.2 kΩ
80
70
RT = 1 kΩ
60
50
RT = 680 Ω
RT = 10 kΩ
40
–55
0.1
10
100
1M
Figure 8
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–35 –15
5
25
45
65
85
TA – Ambient Temperature (°C)
FOSC – Oscillator Frequency (kHz)
Figure 9
68
105
125
Current Mode Controller
AS384x
Typical Performance Curves
Current Sense Input Threshold vs Error Amp Output Voltage
Error Amp Input Voltage vs Ambient Temperature
2.51
1.0
2.50
TA = 25°C
0.8
VFB – Error Amp Input Voltage (V)
VSENSE – Current Sense Input Threshold (V)
1.2
TA = 125°C
0.6
0.4
0.2
TA = –55°C
0
2.49
2.48
2.47
–0.2
VFB = VCOMP
VCC = 15 V
2.46
–60
–0.4
0
1
2
3
4
5
6
–30
0
Figure 10
60
90
120
15
Figure 11
Output Sink Capability in Under-Voltage Mode
Output Saturation Voltage
1A
0
VCC = 6 V
TA = 25°C
–1
VSAT – Output Saturation Voltage (V)
IOUT _ Output Sink Current (mA)
30
TA – Ambient Temperature (°C)
VCOMP – Error Amp Output Voltage (V)
100
10
Source Saturation
VOUT – VCC
TJ = 125°C
–2
TJ = –55°C
3
TJ = 25°C
2
1
Sink Saturation
TJ = 125°C
1
0
0.5
1.0
1.5
2.0
0
10
2.5
VOUT – Output Voltage (V)
Figure 12
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100
IOUT – Output Saturation Current (mA)
Figure 13
69
500
AS384x
Current Mode Controller
Application Information
Section 1 – Theory of Operation
The AS3842/3/4/5 family of current-mode control
ICs are low cost, high performance controllers
which are pin compatible with the industry standard UC3842 series of devices. Suitable for
many switch mode power supply applications,
these ICs have been optimized for use in high
frequency off-line and DC-DC converters. The
AS3842 has been enhanced to provide significantly improved performance, resulting in exceptionally better tolerances in power supply
manufacturing. In addition, all electrical characteristics are guaranteed over the full 0 to 105°C
temperature range. Among the many enhancements are: a precision trimmed 2.5 volt reference
(+/– 1% of nominal at the error amplifier input), a
significantly reduced propagation delay from current sense input to the IC output, a trimmed oscillator for precise duty-cycle clamping, a modified
flip-flop scheme that gives a true 50% duty ratio
clamp on 3844/45 types, and an improved 5 V
regulator for better AC noise immunity. Furthermore, the AS3842 provides guaranteed performance with current sense input below ground.
The advanced oscillator design greatly simplifies
synchronization. The device is more completely
specified to guarantee all parameters that impact
power supply manufacturing tolerances.
The functional block diagram of the AS3842 is
shown in Figure 1. The IC is comprised of the six
basic functions necessary to implement current
mode control; the under-voltage lockout; the reference; the oscillator; the error amplifier; the current
sense comparator/PWM latch; and the output.
The following paragraphs will describe the theory
of operation of each of the functional blocks.
1.1 Under-voltage lockout (UVLO)
The under-voltage lockout function of the
AS3842 holds the IC in a low quiescent current
(≤ 1 mA) “standby” mode until the supply voltage
(VCC) exceeds the upper UVLO threshold voltage. This guarantees that all of the IC’s internal
circuitry are properly biased and fully functional
before the output stage is enabled. Once the IC
turns on, the UVLO threshold shifts to a lower
level (hysteresis) to prevent VCC oscillations.
The low quiescent current standby mode of the
AS3842 allows “bootstrapping”–—a technique
used in off-line converters to start the IC from the
rectified AC line voltage initially, after which power
to the IC is provided by an auxiliary winding off the
power supply’s main transformer. Figure 14 shows
a typical bootstrap circuit where capacitor (C) is
VDC
& gt; 1 mA
R
R & lt;
VDC MIN
AS384x
7
PRI
VCC
1 mA
IC ENABLE
AC LINE
OUT
5
16 V/10 V (3842/4)
GND
+
8.4 V/7.8 V (3843/5)
+
AUX
C
Figure 14. Bootstrap Circuit
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6
70
SEC
AS384x
Current Mode Controller
the following paragraphs. The basic operation of
the oscillator is as follows:
The nature of the AS3842 oscillator circuit is such
that, for a given frequency, many combinations of
RT and CT are possible. However, only one value
of RT will yield the desired maximum duty ratio at
a given frequency. Since a precise maximum
duty ratio clamp is critical for many power supply
designs, the oscillator discharge current is
trimmed in a unique manner which provides significantly improved tolerances as explained later
in this section. In addition, the AS3844/5 options
have an internal flip-flop which effectively blanks
every other output pulse (the oscillator runs at
twice the output frequency), providing an
absolute maximum 50% duty ratio regardless of
discharge time.
A simple RC network is used to program the frequency and the maximum duty ratio of the
AS3842 output. See Figure 15. Timing capacitor
(CT) is charged through timing resistor (RT) from
the fixed 5.0 V at VREG. During the charging time,
the OUT (pin 6) is high. Assuming that the output
is not terminated by the PWM latch, when the
voltage across CT reaches the upper oscillator
trip point (≈3.0 V), an internal current sink from
pin 4 to ground is turned on and discharges CT
towards the lower trip point. During this discharge time, an internal clock pulse blanks the
output to its low state. When the voltage across
CT reaches the lower trip point (≈1.3 V), the current sink is turned off, the output goes high, and
the cycle repeats. Since the output is blanked
during the discharge of CT, it is the discharge
time which controls the output deadtime and
hence, the maximum duty ratio.
1.3.1 Selecting timing components RT
and CT
The values of RT and CT can be determined
mathematically by the following expressions:
D
CT =
R T ƒOSC
K
ln L
KH
=
1.63D
R T ƒOSC
7 VCC
CT
8
5 V REG
OUTPUT
PWM
RT
6 OUTPUT
CLOCK
Large RT/Small CT
4
OSCILLATOR
CT
ID
CT
AS3842
OUTPUT
Small RT/ Large CT
5 GND
Figure 15. Oscillator Set-up and Waveforms
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72
(1)
Current Mode Controller
1
RT
(KL)
V
= REG •
ID
(KL)
D
AS384x
Table 1. RT vs Maximum Duty Ratio
1
– (KH)
D
(2)
1–D
(0.736)
KL =
KH =
VREG − V L
VREG
VREG − VH
VH
D
560
− (0.432)
− (0.432)
683
≈ 0.736
≈ 0.432
81%
85%
88%
90%
3,900
91%
4,700
93%
5,600
94%
6,800
95%
8,200
96%
10,000
97%
18,000
98%
that compensates for all of the tolerances within
the device (such as the tolerances of VREG, propagation delays, the oscillator trip points, etc.)
which have an effect on the frequency and maximum duty ratio. For example, if the combined
tolerances of a particular device are 0.5% above
nominal, then ID is trimmed to 0.5% above nominal. This method of trimming virtually eliminates
the need to trim external oscillator components
during power supply manufacturing. Standard
3842 devices specify or trim only for a specific
value of discharge current. This makes precise
1.3.2 Oscillator enhancements
The AS3842 oscillator is trimmed to provide
guaranteed duty ratio clamping. This means that
the discharge current (ID ) is trimmed to a value
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77%
3,300
Table 1 lists some common values of RT and the
corresponding maximum duty ratio. To select the
timing components; first, use Table 1 or equation
(2) to determine the value of RT that will yield the
desired maximum duty ratio. Then, use equation
(1) to calculate the value of CT. For example, for
a switching frequency of 250 kHz and a maximum duty ratio of 50%, the value of RT, from
Table 1, is 683 Ω. Applying this value to equation
(1) and solving for CT gives a value of 4700 pF. In
practice, some fine tuning of the initial values
may be necessary during design. However, due
to the advanced design of the AS3842 oscillator,
once the final values are determined, they will
yield repeatable results, thus eliminating the
need for additional trimming of the timing components during manufacturing.
72%
2,700
where fosc is the oscillator frequency, D is the
maximum duty ratio, VH is the oscillator’s upper
trip point, VL is the lower trip point, VR is the Reference voltage, ID is the discharge current.
66%
2,200
(4)
63%
1,800
(3)
58%
1,500
D
54%
1,200
1–D
50%
910
1
D
37%
1,000
= 582 •
D
D
22%
820
1
(0.736)
1–D
– (KH)
Dmax
750
D
RT (Ω)
470
1–D
73
AS384x
Current Mode Controller
1.4 Error amplifier (COMP)
and repeatable duty ratio clamping virtually
impossible due to other IC tolerances. The
AS3844/5 provides true 50% duty ratio clamping
by virtue of excluding from its flip-flop scheme,
the normal output blanking associated with the
discharge of CT. Standard 3844/5 devices
include the output blanking associated with the
discharge of CT, resulting in somewhat less than
a 50% duty ratio.
The AS3842 error amplifier is a wide bandwidth,
internally compensated operational amplifier
which provides a high DC open loop gain (90
dB). The input to the amplifier is a PNP differential pair. The non-inverting (+) input is internally
connected to the 2.5 V reference, and the inverting (–) input is available at pin 2 (VFB). The output of the error amplifier consists of an active
pull-down and a 0.8 mA current source pull-up as
shown in Figure 17. This type of output stage
allows easy implementation of soft start, latched
shutdown and reduced current sense clamp
functions. It also permits wire “OR-ing” of the
error amplifier outputs of several 3842s, or complete bypass of the error amplifier when its output
is forced to remain in its “pull-up” condition.
1.3.3 Synchronization
The advanced design of the AS3842 oscillator
simplifies synchronizing the frequency of two or
more devices to each other or to an external
clock. The RT/CT doubles as a synchronization
input which can easily be driven from any open
collector logic output. Figure 16 shows some
simple circuits for implementing synchronization.
8
Open
Collector
Output
5V
VREG
Open
Collector
Output
AS3842
RT
4
RT/CT
3K
CMOS
RT/CT
GND
3K
RT/CT
5
2K
CT
SYNC
2K
EXTERNAL CLOCK
Figure 16. Synchronization
1
VOUT
COMP
COMPENSATION
NETWORK
E/A
2
–
VFB
TO
PWM
+
2.50 V
Figure 17. Error Amplifier Compensation
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0.8 mA
74
Current Mode Controller
AS384x
In most typical power supply designs, the converter’s output voltage is divided down and monitored at the error amplifier’s inverting input, VFB.
A simple resistor divider network is used and is
scaled such that the voltage at VFB is 2.5 V when
the converter’s output is at the desired voltage.
The voltage at VFB is then compared to the internal 2.5 V reference and any slight difference is
amplified by the high gain of the error amplifier.
The resulting error amplifier output is level shifted
by two diode drops and is then divided by three to
provide a 0 to 1 V reference (VE) to one input of
the current sense comparator. The level shifting
reduces the input voltage range of the current
sense input and prevents the output from going
high when the error amplifier output is forced to its
low state. An internal clamp limits VE to 1.0 V. The
purpose of the clamp is discussed in Section 1.5.
and in particular, the characteristics of the major
functional blocks within the supply — i.e. the error
amplifier, the modulator/switching circuit, and the
output filter. In general, the network is designed
such that the converter’s overall gain/phase
response approaches that of a single pole with a
–20 dB/decade rolloff, crossing unity gain at the
highest possible frequency (up to fSW/4) for good
dynamic response, with adequate phase margin
( & gt; 45°) to ensure stability.
Figure 18 shows the Gain/Phase response of the
error amplifier. The unity gain crossing is at
1.2 MHz with approximately 57°C of phase margin. This information is useful in determining the
configuration and characteristics required for the
compensation network.
One of the simplest types of compensation networks is shown in Figure 19. An RC network provides a single pole which is normally set to
compensate for the zero introduced by the output
capacitor’s ESR. The frequency of the pole (fP) is
determined by the formula;
1.4.1 Loop compensation
Loop compensation of a power supply is necessary to ensure stability and provide good line/load
regulation and dynamic response. It is normally
provided by a compensation network connected
between the error amplifier’s output (COMP) and
inverting input as shown in Figure 17. The type of
network used depends on the converter topology
80
ƒP =
1
(5
2π Rƒ Cƒ
240
210
Gain
60
150
40
120
90
20
60
Phase (Degrees)
Gain (dB)
CF
180
Phase
RI
–
E/A
+
RBIAS
30
0
RF
VOUT
To PWM
0
–30
–20
101
2.50 V
–60
102
103
104
105
106
107
Frequency (Hz)
Figure 18. Gain/Phase Response of the AS3842
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Figure 19. A Typical Compensation Network
75
Current Mode Controller
AS384x
AS3842/3/4/5
VIN
COMP
1
VREG
ERROR AMP
8
+
2.5 V
2
VFB
–
PWM
COMPARATOR
VE
–
1V
3
R
CURRENT
SENSE
RT/CT
4
PRI
SEC
7
PWM LOGIC
FF
S
R
+
VCC
5 V REG
2R
CLOCK
OUTPUT
6
GND
CLOCK
VE
VS
5
OUTPUT
IS
VS
R
C
RS
Leading Edge Filter
Figure 20. Current Sense/PWM Latch Circuit and Waveforms
1.6 Output (OUT)
The current sense comparator’s inverting input
is internally clamped to a level of 1.0 V to
provide a current limit (or power limit for multiple
output supplies) function. The value of RS is
selected to produce 1.0 V at the maximum
allowed current. For example, if 1.5 A is the
maximum allowed peak inductor current, then
RS is selected to equal 1 V/1.5 A = 0.66 Ω. In
high power applications, power dissipation in
the current sense resistor may become
intolerable. In such a case, a current
transformer can be used to step down the
current seen by the sense resistor. See
Figure 21.
The output stage of the AS3842 is a high current
totem-pole configuration that is well suited for
directly driving power MOSFETs. It is capable of
sourcing and sinking up to 1 A of peak current.
Cross conduction losses in the output stage have
been minimized resulting in lower power dissipation in the device. This is particularly important
for high frequency operation. During undervoltage shutdown conditions, the output is active
low. This eliminates the need for an external pulldown resistor.
1.7 Over-temperature shutdown
The AS3842 has a built-in over-temperature
shutdown which will limit the die temperature to
130°C typically. When the over-temperature condition is reached, the oscillator is disabled. All
other circuit blocks remain operational. Therefore, when the oscillator stops running, output
pulses terminate without losing control of the
supply or losing any peripheral functions that
may be running off the 5 V regulator. The output
may go high during the final cycle, but the PWM
N:1
VS
VS =
IS
N
RS
RS
IS
Figure 21. Optional Current Transformer
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77
AS384x
Current Mode Controller
latch is still fully operative, and the normal termination of this cycle by the current sense comparator will latch the output low until the
over-temperature condition is rectified. Cycling
the power will reset the over-temperature disable
mechanism, or the chip will re-start after cooling
through a nominal hysteresis band.
A simple RC filter is used to suppress the spike.
The time constant should be chosen such that it
approximately equals the duration of the spike. A
good choice for R1 is 1 kΩ, as this value is optimum for the filter and at the same time, it simplifies the determination of RSLOPE (Section 2.2). If
the duration of the spike is, for example, 100 ns,
then C is determined by:
Section 2 – Design Considerations
2.1 Leading edge filter
C =
The current sensed by RS contains a leading
edge spike as shown in Figure 20. This spike is
caused by parasitic elements within the circuit
including the interwinding capacitance of the
power transformer and the recovery characteristics of the rectifier diode(s). The spike, if not properly filtered, can cause stability problems by
prematurely terminating the output pulse.
Ve
=
Time Constant
(6)
1 kΩ
100 ns
1 kΩ
= 100 pF
2.2 Slope compensation
Current-mode controlled converters can experience instabilities or subharmonic oscillations
Ve
IPK
IAVG 2
∆I
IL2
∆I'
IAVG 1
m
2
1
m
1
m
m
2
IL1
T0
D1
D2
T1
T0
D1
(a)
D2
T1
(b)
VCOMP
VCOMP
m=m
m=m
2 /2
m
1
m
1
2 /2
IL2
IL1
T0
∆I
m
IAVG 1 = IAVG 2
D1
D2
2
T1
T0
(c)
D1
D2
(d)
Figure 22. Slope Compensation
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∆I'
m
2
78
T1
Current Mode Controller
AS384x
the oscillator at pin 4, it is more practical to add
the slope compensation to the current waveform.
This can be implemented quite simply with the
addition of a single resistor, RSLOPE, between pin
4 and pin 3 as shown in Figure 23(a). RSLOPE, in
conjunction with the leading edge filter resistor,
R1 (Section 2.1), forms a divider network which
determines the amount of slope added to the
waveform. The amount of slope added to the current waveform is inversely proportional to the
value of RSLOPE. It has been determined that the
amount of slope (m) required is equal to or
greater than 1/2 the downslope (m2) of the inductor current. Mathematically stated:
when operated at duty ratios greater than 50%.
Two different phenomena can occur as shown
graphically in Figure 22.
First, current-mode controllers detect and control
the peak inductor current, whereas the converter’s output corresponds to the average inductor current. Figure 22(a) clearly shows that the
average inductor current (I1 & I2) changes as the
duty ratio (D1 & D2) changes. Note that for a fixed
control voltage, the peak current is the same for
any duty ratio. The difference between the peak
and average currents represents an error which
causes the converter to deviate from true
current-mode control.
Second, Figure 22(b) depicts how a small perturbation of the inductor current (∆I) can result in an
unstable condition. For duty ratios less than 50%,
the disturbance will quickly converge to a steady
state condition. For duty ratios greater than 50%,
∆I progressively increases on each cycle, causing an unstable condition.
m ≥
Slope compensation can also be used to improve
noise immunity in current mode converters operating at less than 50% duty ratio. Power supplies
operating under very light load can experience
8
VREG
4
4
RT/CT
OPTIONAL
BUFFER
AS3842
IS
CT
R1
IS
3
AS3842
R1
3
ISENSE
ISENSE
GND
RS
5
(a)
5
(b)
Figure 23. Slope Compensation
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RT/CT
CT
RSLOPE
GND
RS
VREG
RT
RT
RSLOPE
(7)
2
In some cases the required value of RSLOPE may
be low enough to affect the oscillator circuit and
thus cause the frequency to shift. An emitter follower circuit can be used as a buffer for RSLOPE
as depicted in Figure 23(b).
Both of these problems are corrected simultaneously by injecting a compensating ramp into
either the control voltage (VE) as shown in Figure
22(c) & (d), or to the current sense waveform at
pin 3. Since VE is not directly accessible, and, a
positive ramp waveform is readily available from
8
m2
79